FIG. 6 is a block circuit diagram of a conventional electric power converter. The conventional electric power converter has a circuit configuration similar to that of the switching power supply disclosed in Unexamined Japanese Patent Application 2004-153948, equivalent to U.S. Pat. No. 6,917,528 B2. In the circuit shown in FIG. 6, a main switching device 1 and a subsidiary switching device 2 repeat switching ON and OFF alternately such that the excitation energy stored in a transformer 6 while main switching device 1 is turned ON is fed to a load as a DC output while main switching device 1 is turned OFF.
Now the operations of the circuit shown in FIG. 6 will be described with reference to FIG. 7. FIG. 7 is a wave chart describing the operations of the circuit shown in FIG. 6. In the following descriptions, the main and subsidiary switching devices are metal oxide semiconductor field-effect transistors (MOSFETs).
Referring now to FIG. 7, the voltage between the gate and source (hereinafter referred to as the “gate-source voltage”) VGS1, the voltage between the drain and source (hereinafter referred to as the “drain-source voltage”) VDS1 and the drain current ID1 of main switching device 1 shown in FIG. 6 are described. Also described in FIG. 7 are the gate-source voltage VGS2, the drain-source voltage VDS2 and the drain current ID2 of subsidiary switching device 2 shown in FIG. 6. The current IDr of a diode 8 in FIG. 6 is described in FIG. 7. As described in FIG. 7, the operations of the circuit shown in FIG. 6 may be considered in time periods from time t1 to time t6.
State 1: t1 to t2
As the gate-source voltage VGS1 of main switching device 1 exceeds a gate threshold voltage VGS(th) to the higher side in the state, in which the gate input capacitance of main switching device 1 is charged up via a resistor 18 by the voltage generated in the third winding 6f of transformer 6 and the body diode in main switching device 1 is electrically conductive such that the drain-source voltage VDS1 is 0, main switching device 1 performs zero-voltage turn-ON in the state, in which a current is flowing through the body diode thereof. The drain current ID1 of main switching device 1 is equal to the exciting current of transformer 6 and increases linearly. Since the gate-source voltage VGS2 of subsidiary switching device 2 is negative due to the voltage generated in the fourth winding 6b of transformer 6, subsidiary switching device 2 is OFF.
State 2: t2 to t3
As the voltage generated across a resistor 17 by the drain current ID1 of main switching device 1 exceeds the voltage between the base and emitter of a transistor 21, transistor 21 turns ON. Since the gate input capacitance of main switching device 1 is discharged, main switching device 1 turns OFF, the drain-source voltage VDS1 of main switching device 1 rises and the drain-source voltage VDS2 of subsidiary switching device 2 lowers.
State 3: t3 to t4
Diode 8 becomes electrically conductive and the excitation energy stored in transformer 6 is fed to the secondary side thereof. Subsequently, the voltage across transformer fourth winding 6b rises and shifts from negative to positive.
State 4: t4 to t40
As the voltage across transformer winding 6b exceeds the gate threshold voltage VGS(th) of subsidiary switching device 2 to the higher side, subsidiary switching device 2 performs zero-voltage turn-ON in the state, in which a current is flowing through the body diode thereof.
State 5: t40 to t5
As all the excitation energy stored in transformer 6 is discharged, diode 8 becomes OFF and the voltage across transformer fourth winding 6b starts lowering.
State 6: t5 to t6
As the voltage across transformer fourth winding 6b falls below the gate threshold voltage VGS(th) of subsidiary switching device 2, subsidiary switching device 2 turns OFF. The drain-source voltage VDS2 of subsidiary switching device 2 rises and the drain-source voltage VDS1 of main switching device 1 lowers.
State 7: from t6
The drain-source voltage VDS1 of main switching device 1 is set at zero and the drain-source voltage VDS2 of subsidiary switching device 2 at the voltage of a DC power supply 3.
The subsequent operations return to those in the State 1 and repeat self-oscillations repeating the States 1 through 6.
Since the main and subsidiary switching devices in the circuit shown in FIG. 6 perform zero-voltage turn-ON, turn-on losses are not caused. Since the magnetic energies stored in the leakage inductances of transformer 6 and reactor 5 are regenerated to the DC power supply or the transformer secondary side, an electric power converter that causes low losses and exhibits a high conversion efficiency is obtained.
Although any control circuit is not disposed for the subsidiary switching device in FIG. 6, an electric power converter including a control circuit, which includes a transistor and a delay circuit, for controlling the subsidiary switching device is described in Unexamined Japanese Patent Application 2002-112544, equivalent to U.S. Pat. No. 6,469,913 B2
It is necessary to design the circuit, which drives a subsidiary switching device with the auxiliary winding (fourth winding 6b) of a transformer as described above, so that the voltage applied between the gate and source of the subsidiary switching device may not exceed the gate breakdown voltage. If the subsidiary switching device is a MOSFET, the gate breakdown voltage will usually be around ±30 V.
Immediately after main switching device 1 starts switching by DC power supply 3, the voltage across a capacitor 4 is zero. Therefore, the maximum value VGSmax of the gate-source voltage of subsidiary switching device 2 is given by the following formula (I).VGSmax=(Voltage of DC power supply 3)×(Number of turns in fourth winding 6b)÷(Number of turns in primary winding 6a)  (1)
Depending on the ON-duty of main switching device 1, the voltage described by the formula (1) may be applied between the gate and source of subsidiary switching device 2.
For example, when DC power supply 3 is obtained by rectifying the commercial AC power supply, the commercial AC power supply voltage is different from country to country. Therefore, if one wants to obtain a switching power supply employable in all the countries, the range of the voltage across DC power supply 3 will be inevitably wide. As the foregoing formula (1) clearly indicates, it is very difficult to design such that the maximum gate-source voltage VGSmax never exceeds the gate breakdown voltage over the entire voltage range of DC power supply 3.
For obviating the problem described above, a Zener diode may be connected between the gate and source of subsidiary switching device 2 to clamp the gate-source voltage of subsidiary switching device 2 with the Zener voltage and a Zener current may be made to flow via a resistor 16 connected to the gate terminal of subsidiary switching device 2, when the gate-source voltage of subsidiary switching device 2 is clamped. However, since subsidiary switching device 2 causes more switching losses as the resistance value of resistor 16 is higher, the resistance value of resistor 16 is set usually at several tens to several hundreds ohm. Therefore, a high Zener current is caused when the gate-source voltage is clamped, further causing a low conversion efficiency.
In view of the foregoing, it would be desirable to provide an electric power converter that facilitates controlling the control terminal voltage (gate voltage) of a subsidiary switching device to be lower than the gate breakdown voltage in a wide DC input voltage range or in various operation modes, reducing the losses thereof and obtaining a high conversion efficiency.